Introduction

I remember listening to my dad's record collection after school (Argent, Foghat, Zepplin,...) and that it was singular task..I wanted to listen to music, not do something else and listen to music..it wasn't about having music in the background.. the music was front and center. The same goes today, when I listen to music I listen to music! I remember it like yesterday when I bought my first album as a teenager..the year was 1980 and the album was Van Halen's Women and Children First. I also remember the year, maybe 1985 or 1986, when Tower Records on Columbus Avenue in San Francisco discontinued records to make way for the future..the Compact Disc (CD). I must have bought over 100 records that summer! Eventually, I ended up buying a CD player, a Luxman D-111 in 1988 and converted to the "dark side" I also remember the day I sold my record collection. Too much to deal with since I was moving around a lot back then. Flash forward to 2017... I have come full circle! Enough nostalgia! Nothing is permanent.

 

Since I have "rediscovered" vinyl and I wanted to design and build a phono pre-amp that I could use with my old-school turntable. I remember how good a moving coil cartridge sounded on my Dual CS5000, so I wanted the phono pre-amp to be a dedicated moving coil unit and I also wanted only two stages of amplification to keep the overall design somewhat "streamlined". This project has been in the works for over a year! It took a lot of research  to get to this point. Below is the schematic I settled for, which is at about at the 95% design stage. There are minor points, such as final values that need to be determined for resistors and capacitors, particularly in the RIAA equalization network; but overall, this is the circuit. If you have viewed my other projects, you will see familiar patterns, such as the use of a CCS for the 5687 triode stage, the use of the LR8 for voltage regulation for both the 6C6 and 5687 stages, and the use of the TL431 for shunt regulation in both stages, as well. The amp is in progress and should be completed by the end of 2017 or early 2018.

Here is the schematic for the phono pre-amp (one-channel shown)

The Circuit 

Before I jump into the discussion of the circuit, I wanted to highlight two projects that I have relied heavily on for design guidance: Pete Millet's Phono Pentode Amp and Bruce Heran's The Groovewatt. The circuit is composed of four stages (1) the moving coil step-up transformer at the input, (2) the 6C6 pentode gain stage, (3) the passive RIAA equalization network, and (4) the 5687 triode gain stage at the output. Each stage is discussed below.

Step-Up Transformer (SUT) Stage

Many moving coil (MC) cartridges have outputs in the ~0.25mV to 1mV range, so a phono pre-amp has to amplify this signal to a level that will provide sufficient drive to a pre-amp or power amp. You can certainly  bypass the SUT and directly amplify the signal in a tube gain stage, but there are several issues to consider. The first is noise. Apparently, it is difficult to amplify low level signals with a tube gain stage without adding audible noise, but trying to design such a tube gain stage could be an interesting learning experience. However, I decided that my effort would be better spent elsewhere since the use of a SUT allows amplification of a low-level signal without adding noise and allows for fine-tuning of the load impedance for the MC cartridge. I opted to use the Lundahl LL9226 MC step-up transformer and wire it for a gain of 20 (26dB). I installed them on a PCB from K & K Audio. Most MC cartridge manufacturers provide a recommended range for load impedance, but I have read that a general rule of thumb is that the load impedance should be ~10 times the output or internal impedance of the MC cartridge. In my design I am using a rotary switch that will allow six settings for load impedance from ~22 ohms to 117 ohms. An important consideration is that with a 47K grid resisitor, the maximum load impedance for the cartridge is 117 ohms because the load reflected from the secondary to the primary of the SUT is 47K/400 = 117.5 ohms. To get a lower load impedance you need to add a resistor in parallel with the 47K resistor; for example, the minimum load impdance setting is 1/ (1/47K + 1/10.7K) = 9.1K/400 = ~22 ohms. The "400" is the impedance ratio of the Lundahl 9226 when wired for 26dB of gain. The impedance transformation through a transformer is equal to the square of the voltage gain, so in this example, the voltage gain is 20, so 20 x 20 = 400. If you wanted to increase the load impedance range, you would have to increase the value of the grid resistor or reduce the gain of the SUT. Here is a link for a useful calculator for determining load impedance and a link on transformer basics.  The amp should work well for MC cartridges with an output impedance < 10 ohms.  The overall gain of the SUT stage is 20, so a 0.5mV input signal would be amplified to 10mV, which is a level that should give an acceptable signal to noise ratio(S/N) for the pentode gain stage.

The Pentode Gain Stage

The great thing about a pentode gain stage is that you get a lot of gain in a single stage. However, there is no free lunch because all that gain comes with several issues that need to be addressed. Here is a brief overview of pentodes. In general, a pentode is on average, noisier than a triode. The noise is inherent to how the pentode operates and is generally classified as "tube noise". However, the magnitude of this noise will vary from individual tube as well as type of tube. Here is an interesting thread on pentode noise.  Anyway, "tube noise" does not bother me that much as long as it does not intrude on the music. In my opinion, I think input noise and power supply ripple are bigger concerns. The pentode I decided on is the 6C6. You might be wondering if there is anything special about the 6C6? The answer is sort of no, except that the 6C6 is very similar electrically to the more well known 6J7 but the 6C6 uses a small 6-pin base. It has been used as a driver for the 2A3 and 300B DHT tubes in some well regarded SE amps. I also have a bunch of 6C6 tubes. The gain of the pentode stage is ~100 (plate resistor x transconductance or 82K x 1.185), so noise on the input needs to be minimized. As mentioned above, the  use of the SUT brings the input signal up to 10mV from a 0.5mV MC cartridge output without appreciable noise. Other design strategies include the use of shielded signal wire from the amp inputs to the SUT and from the SUT to the grid cap, the use of grid stopper resistor, and of course minimizing noise from the turntable (a subject for another section). I use Beldon 8450 shielded wire and find it works well and easy to work with. The 6C6 also has an internal shield at the top of the tube near the top cap attachment for the control grid. However, this shield might be there to prevent oscillation when the tube is used for radio frequency detection and amplification.

 

Pentodes have a poor Power Supply Rejection Ratio (PSRR), so an extremely quiet power supply needs to be used to minimize ripple on the output (hum). In this circuit the PSRR of the 6C6 gain stage is ~ 1! ((82K + 1M)/1M)). The operating point of the pentode is 2mA of plate current and 0.5mA of screen grid current. The plate voltage is ~100-105V and the supply voltage is 270V. The grid to cathode voltage is 3V. The supply voltage is regulated by a LR8 regulator. The screen voltage is regulated at 100V by an LR8 regulator, as well. The LR8s provide 60dB of ripple rejection and have proved excellent in my other projects. Read more here and here. To reduce the output noise level of the LR8, which is already low, I added a 1uF capacitor from the voltage divider junction to ground.  Another important consideration with pentodes is that you want to make sure there is no or a very low AC impedance between the cathode and the screen grid or a AC signal could appear on the screen grid resulting in negative feedback, which would reduce gain. The screen grid circuit uses a 10uF low-impedance electrolytic and a 0.1uF film capacitor, which should pass all audio frequencies. The calculated ripple voltage on the plate and screen grid is ~0.06uV RMS. The RC (resistor-capacitor) filter stage gives a ripple rejection value of ~67 and the LR8 regulator gives a ripple rejection value of ~1000, so the 4mV ripple from the power supply (see Power Supply section below) would be attenuated by 4mV/67 = 0.06mV/1000 = .00006mV (~.0.06uV).

I also wanted to dial in the operating point of the 6C6, so I needed the ability to adjust the grid to cathode voltage.  Texas Instruments documents a circuit using the TL431 regulator that I used here. The TL431 adjustable shunt regulator works very well and you can adjust the bias of the cathode by simply replacing one of the programming resistors with a trimmer resistor. The only concern is that you can not set the voltage below 2.5 V, which is the reference voltage of the TL431. The range of adjustment is ~2.5 V to 15 V.  Since the circuit uses the TL431 in the TO-92 package, I like to keep power dissipation less than 0.25 W, which is not an issue for the operating point of the 6C6. Noise output is ~10uV, so very low. The TL431 has a low dynamic impedance, so no bypass capacitor is needed to realize full gain. The output impedance of the tube is essentially equal to the anode resistor value.

The RIAA Stage

The RIAA stage is a passive circuit as opposed to an active RIAA circuit. Here is a good discussion on RIAA equalization. Since this is my first phono pre-amp, I decided to keep the RIAA circuit as simple as possible and passive RIAA circuits are pretty simple, at least on paper. I used this handy calculator to calculate the values of the capacitors and resistor (s) needed. The values are dependent of the output impedance of the pentode gain stage, which is essentially the value of the plate resistor, which is 82K. The nice thing about pentodes is that the plate resistance (~1M for 6C6) is much larger than the plate resistor (82K) so the plate resistance has very little effect on the source impedance driving the RIAA stage, at least on paper. If you do the calculations, you arrive at a source impedance of  ~76K (1/(1/1M + 1/.082K). This is the impedance value I used in calculating the capacitor and resistor values for the RIAA network. Luckily, the calculated values are very close to readily available values. If not, series resistance can be added to adjust the source impedance. The other thing to consider is to make sure that the coupling capacitor that passes the AC signal to the 5687 stage is sufficiently large to pass all audio frequencies. You can use this calculator as a guide. I opted to start with a 0.047uF capacitor and go from there. The -3dB frequency response is 3.39 Hz and the optimal low frequency response is ~ 34Hz. After the amp is up and running, I will likely need to verify the accuracy of RIAA network and make adjustments, if necessary. One last thing, the RIAA network attenuates the AC signal so there is a loss of gain of ~10. If the output from the pentode gain stage is ~1.0V (assuming a 10 mV input), then the input voltage to the 5687 stage would be ~100mV.

The 5687 Triode Gain Stage

The goal of the triode gain stage is to compensate for the loss of gain in the RIAA network and provide a relatively low source impedance for the next stage, which in this case will be my DHT pre-amp. I became familiar with the dual-triode 5687 tube several years ago when I built a headphone amp using this tube. The 9-pin 5687 is a punchy little tube with incredible imaging and a plate resistance of only~2500 ohms, which means is can easily drive the 100K input impedance of my DHT pre-amp. Gain is ~17, which is little higher than needed but will work fine. The overall gain of the amp will be ~3400 or 70B, so 0.5mV at the input will give a ~1.7V at the output. The 5687 supply voltage is regulated by a LR8 regulator and a CCS plate load. The benefits of using a LR8 are discussed above in the Pentode Gain Stage. Also, since it is a dual triode, only one tube is needed in the phono pre-amp.

There is a lot of information on why constant current sources (CCS) are a good way to go for loading the plate of a triode, so I will not go into detail of the theory on how they work and why they perform so well. Read more here. Once again, no reason to reinvent the wheel since there are many great CCS circuits that have been developed. An excellent reference for CCS circuits  can be found in the book Valve Amplifiers by Morgan Jones. I opted to use a cascode CCS circuit similar to Bottlehead's C4S design, but I added a trimmer resistor that allows for adjustment of the current through the CCS as well as a proper heatsink. The 47.5 ohm resistor in series with the trimmer resistor is there just so there is an upper limit to how much current can flow through the CCS.  The current is adjustable from around 6.5 mA to 19.5 mA by varying the resistance with the trimmer resistor (the lower the resistance the greater the current). The nice thing about using a trimmer resistor is that you can just dial in the current. The MJE5731 gets a heatsink since it drops most of the voltage, which is ~144 volts and dissipates ~2.2 watts of power. Another thing to consider is the power rating for the LR8 voltage regulator in the TO-92 package , which is only 0.74 watts. I like to keep the power dissipation below 0.4 watts just to keep the regulator well within its operating parameters. In this case, the LR8 dissipates ~0.43W, which is probably OK. I definitely would not want the LR8 to dissipate any more power. Be sure to factor in the reference currents used by the LR8 (1mA) and the CCS circuit (1mA) when computing power dissipation. A CCS circuit allows one to easily change the operating conditions of a triode and reduces distortion; therefore, it made sense to implement it in the design of this pre-amp. Another benefit of a CCS over an anode load resistor in a cathode bias gain stage is a ~10-fold improvement in PSSR (power supply rejection ratio or 40dB).  The calculated ripple voltage at the output of the 5687 tube is ~0.04uV RMS (4mV from power supply/1000 =0.004mV/100 = 0.04uV).

Again, since I wanted to dial in the operating point of the 5687 I needed to adjust the grid to cathode voltage. Texas Instruments documents a circuit using the TL431 regulator that I used for this pre-amp here. The TL431 adjustable shunt regulator works very well and you can adjust the bias of the cathode by simply replacing one of the programming resistors with a trimmer resistor. The only concern is that you can not set the voltage below 2.5 V, which is the reference voltage of the TL431. The range of adjustment is ~2.5 V to 15 V.  Since the circuit uses the TL431 in the TO-92 package, I like to keep power dissipation less than 0.25 W, which is not an issue for the operating point of the 5687. Noise output is ~10uV, so very low. The TL431 has a low dynamic impedance, so no bypass capacitor is needed to realize full gain. Even with no bypass capacitor the output impedance of the tube is essentially equal to the plate resistance of that tube, which will vary slightly with plate current. As far as noise or sound quality goes, I can not hear any difference between a resistor bypassed with a capacitor and the TL431 regulator. The output capacitor is 1.0uF, which will give optimal frequency response  down to 16Hz with a 100K load impedance. I did not include an output load resistor in the pre-amp since it is used to drive pre-amps or power amps with a source impedance of at least 100K. This pre-amp could probably handle pre or power amps with source impedances down to 25K since the output impedance of the circuit is only ~2500 ohms, but the output capacitor value would need to be increased to maintain low end frequency response. Some builders use a series resistor in the output to maintain a linear response for long cables, but I have never had issues not using them and I use cables up to 5 feet. Also, the output signal from the phono pre-amp will go into my DHT pre-amp through ~3 foot cables.

The Power Supply

Here is a picture of the Glassware PS-1 Power Supply and Schaffner EMI-filtered Power Entry Module (FN283-02-06):

Here is a picture of the Lundahl SUT transformers, Neutrik locking 1/4" chassis jacks used for the input to the grid of the 6C6, the Grayhill Series 44 6-position, 2-pole, 1-deck rotary switch (44D30-01-2-ADN) for adjusting the MC cartridge load impedance, and the input and output RCA jacks. The 9-pin socket is under the SUT PCB board. The tall stand-offs are for the PCB boards for each channel. Once I have the PCBs installed, I will update the website. There is also a shield between the power supply section and the rest of the circuit to help reduce potential power supply noise from getting into the pentode gain stages:

The power supply uses a stock Glassware Audio Design PS-1 except for IXYS DSEI-12A fast recovery rectifiers, MBR735G Schottky rectifiers, and Nichicon KX 220uF, 400V audio-grade capacitors. The PS-1 PCB has a circuit with a high voltage current regulator paired with a low-voltage regulator, which supplies the B+ and a low voltage regulated DC circuit which supplies the tube filaments. You can read more about the PS-1 here. I have used the PS-1 in other projects with great results and highly recommend it. The output voltage is set at 300 volts by two 30K resistors that form a voltage divider (see schematic below). The transformer is made by Edcor (XPWR193) and the two secondaries are rated at 240V at 200mA and 9V at 4A. Edcor transformers are excellent and probably the best value out there. They are made to order, so you have to wait several weeks but they have always delivered my orders early. The total current draw of the amp is ~50mA. The rectified voltage is ~330V, so the IXCY 10M45S drops ~28V and dissipates a reasonable ~1.4 watts of power. Here is a handy design guide to rectifier use. The power supply is reasonable quiet with a calculated output ripple of ~4mV RMS into the LR8 regulator that feeds the CCS of the 5687 triode gain stage and the RC stage that feeds the LR8 regulators that in turn feed the screen gride and plate of the 6C6 pentode gain stage. 

I like to pretty much use regulated DC for filament heating in all my projects to eliminate or minimize hum. The filaments of the 5687 can be wired in series at 12.6V at 0.45 amps (pins 4 and 5 connected to + and - of the filament regulator output, respectively) or in parallel at 6.3V at 0.9 amps (pin 4 and 5 connected to the + output, and pin 8 connected to the - output). The filament of the 6C6 is rated at 6.3V at 0.3A. Since I wanted to use one filament regulator for the amp, the output voltage was set at 6.3V to power the filaments for both the 5687 and 6C6 tubes. The LD1085 needs a minimum input-output voltage of 1.3V for it to regulate, but I like to leave a bit of headroom by keeping the voltage differential above 2.5V.  I wired the 9V secondary to a full-wave bridge rectifier, which gives a voltage input to the regulator of ~11.5V leaving ~5.2V, which is way more than needed!. The power dissipation through  the regulator is ~7.8 watts, which is on the high side. Hopefully, the heat sink is up to the task with a thermal resistance of 7 degrees Celsius per watt. This would be marginal if the ambient temperature approaches ~100 °F. An option with the PS-1 board is to bias the filament output with a voltage divider from the B+, which is highly recommended since this tends to reduce noise. For this pre-amp, I set the bias at ~75V (the heater-cathode voltage difference cannot exceed 90V DC). Noise output is ~ 19mV RMS, which is fine for an indirectly heated cathode. The schematic is below. 

Here is the schematic of the PS-1 power supply:

Here is the PS-1 filament circuit: